Power components are defined below as elements which are used, for example, in automotive and industrial electronics and have at least one component provided for the switching of voltages and currents. The voltage and current range of such power or high-power components is in a range of 5 V to 6500 V per component and current intensities of a few milliamperes to hundreds of amperes per component.
The present invention relates to an insulated gate bipolar transistor (IGBT), which usually at least includes a zone of a second conduction type (collector layer), a region (drift zone) of the first conduction type, a body region of the second conduction type, into which well zones of the first conduction type are embedded, and a gate electrode above the well zone, the gate electrode being spaced apart by means of an insulating layer. Here and hereinafter an IGBT shall also be understood to mean an IEGT (IEGT=Injection Enhanced Gates Transistor).
The present invention relates in particular to a trench IGBT that is used at higher voltages from approximately 600 V upward.
The basic structure of such an IGBT is illustrated in FIG. 1a. 
A semiconductor body has a p-conducting collector zone 1 provided with a metallization 13, on which collector zone a first, n-conducting body zone (drift zone) 2 and a second, p-conducting body zone 3 are provided successively. An n-conducting emitter zone 4 is embedded into the p-conducting body zone 3. IGBT cells are illustrated in the present example, so that two emitter zones 4 are correspondingly present.
Trenches 5, 6 penetrate through the emitter zone 4 and the p-conducting body zone 3, and reach right into the n-conducting body zone. Said trenches 5, 6 are lined with an insulating layer 7 made of silicon dioxide for example, the insulating layer acting as gate oxide. The interior of the trenches 5, 6 is filled with a polycrystalline silicon that forms a gate electrode 8. The gate electrode 8 is covered with an insulating layer 10 made of, for example, silicon dioxide, silicon nitride or made of amorphous silicon carbide, with the result that the conductive material 8 is electrically isolated from a metallization serving as emitter contact 11 on a main surface 9.
Dopings can be configured oppositely in each case, so that the p-type collector layer 1 designated for example in FIG. 1a can also become an n-type collector layer, in this case the drift zone 2 having p-doping, the body zone 3 being n-doped and the wells 4 incorporated into the body zone 3 being p-doped.
As is illustrated in FIG. 1b, a semiconductor component includes an insulated gate trench IGBT including many, generally many hundreds of, IGBT cells connected in parallel. One cell is illustrated in FIG. 1a, while FIG. 1b represents a plurality of such cells in plan view which are arranged in a cell array. IGBTs and IEGTs are particularly suitable for higher voltages, as has already been mentioned, and three examples of said IGBTs and IEGTs are shown in FIGS. 2, 3, and 4.
In the example of FIG. 2, the distance between two cells is extended by an optional additional trench 14 without an emitter zone, while in the examples of FIGS. 3 and 4 a relatively wide p-conducting zone 15 (p-float) is located between the corresponding two cells, which zone may overlap the trench edge with a bulge 17. In the example of FIG. 4, additionally relative to the example of FIG. 3, the conductive materials 8 in the trenches 5, 6 of the adjacent cells are also connected to one another by means of a conductor layer 16. The insulating layer 7 may, if appropriate, also be made thicker at the side remote from the emitter zone 4.
High-voltage switches in power electronics, such as IGBTs and the like, are usually designed in such a way that their reverse voltage lies significantly above the typical operating voltage predefined by the application. If the operating voltage lies within the range of 600-850 V, for example, then a switch module (e.g. IGBT) made available for this is designed for example in such a way that it has a rated reverse voltage of 1200 V. As has been illustrated, such a large safety margin is necessary in order to absorb the high overvoltages that occur when the high-voltage switches are turned off, in order that destruction of the IGBT is reliably avoided. The high overvoltages are caused by high rates of current change di/dt and the always present (including parasitic) inductances (“leakage inductances”). Particularly high overvoltages occur primarily when a multiple of the rated current is to be switched off, as in the so-called overcurrent or short-circuit case.
In this regard, primarily the modern high-voltage switches which switch (off) very rapidly have proved to be problematic, which switches tend toward very high overvoltages at high leakage inductances. Examples of such modern rapidly switching IGBTs are described in T. Laska et al. “The Field Stop IGBT (FS IGBT)—A New Power Device Concept with a Great Improvement Potential”, Proceedings of the 12th ISPSD, pages 355-358, 2000; M. Otsuki et al. “Investigation of the Short-Circuit Capability of 1200V Trench Gate Field-Stop IGBTs, Proceedings of the 14th ISPSD, pages 281-284, 2002; S. Dewar et al. “Soft Punch Through (SPT)—Setting New Standards in 1200V IGBT”, Proc. PCIM Europe, 2000; K. Nakamura et al. “Advanced Wide Cell Pitch CSTBTs Having Light Punch-Through (LPT) Structures”, Proceedings of the 14th ISPSD, pages 277-280, 2002. An FS-IGBT differs from the IGBT illustrated in FIG. 1 by virtue of the fact that a more highly defined n-type field stop layer is arranged between the p-conducting region 1 and the n-conducting layer of the base or drift zone 2. Moreover, further variations of an IGBT are known, such as e.g. a punch-through IGBT (PT-IGBT), which involve the drift zone being subdivided into a plurality of differently doped n-type regions.
FIG. 5 represents a V/t diagram in which the voltages V that occur during switch-off are plotted as a function of the time t. As can be gathered from FIG. 5, during the switching operation of a conventional IGBT, without regulating measures, the overvoltage would rise above the breakdown voltage (cf. for example “1200 V”) and possibly destroy the component (curve a). When using conventional IGBTs, therefore, it is necessary to comply with a sufficient safety margin between the intermediate circuit voltage and the static breakdown voltage of the component.
In order that the overvoltage is always kept below the breakdown voltages, the switching speed is reduced in the conventional IGBTs to an extent such that the overvoltage generated at leakage inductances reliably remains below the specific breakdown voltage (e.g. 1200 V) in every conceivable operating and disturbance case. A curve b shows the behavior of an IGBT in which the switching speed has been reduced to an extent such that the breakdown voltage of 1200 V is not reached. However, high switching losses are accepted in the case of this concept.
One possibility for minimizing the switching losses consists in limiting the overvoltage by means of active intervention in the switching operation with e.g. reactivation of the switch or “clamping” with zener diodes. This switching operation is represented by a curve c in FIG. 5. However, the concept used in this case requires an additional outlay on the driving and circuitry. In both cases, the specified breakdown voltage (e.g. 1200 V) lies significantly above the usable intermediate circuit voltage range of e.g. 650-850 V.
The components are ideally intended to have the property of dynamically limiting the overvoltage. In this case, the level of the dynamic limiting voltage should lie in the region of the previously used and specified breakdown voltage of 1200 V, for example, in order to achieve low switching losses. The dynamic limiting would have the advantage that the static breakdown voltage of the components can lie below the voltage in this case. The static breakdown voltage is in this case intended to lie scarcely above the maximum intermediate circuit voltage of 1000 V, for example. The voltages of such a component which occur during the switching operation are shown by a curve d in FIG. 5. Such a component could be designed for a low breakdown voltage. Both the on-state losses and switching losses can be reduced by means of the small thickness of the component. In operating cases in which the component dynamically limits its overvoltage, the minimum possible switching losses for this case (leakage inductance and current intensity) arise.
This possibility of limiting overvoltages in IGBTs has recently been discussed as the dynamic “self-clamping property” (dynamic clamping) of an IGBT, which is based on the changes in the net space charge density of the drift region on account of an avalanche current, depleted charge carriers of the neutral region and the collector-side injection of holes (see e.g. B. M. Takei et al. “Analysis on the Self-Clamp Phenomena of IGBTs” in Proc. 11th ISPSD, Poster Session, Paper 7.1, 1999 and M. Otsuka “1200V FS-IGBT module with enhanced dynamic clamping capability”, Proceedings of the 2004 ISPSD, pages 339-342, 2004).
If it is accordingly possible to configure a high-voltage switch such that, despite the presence of high leakage inductances and high rates of current change, it is able to dynamically limit a high overvoltage occurring during turn-off by means of self-clamping (“dynamic clamping”) and, in the case of this loading (which may amount to as much as a multiple of the rated current at the rated voltage, for instance), upon the overshooting of a critical reverse voltage (clamping voltage), to have the ability to carry a high current of the order of magnitude of the rated current and above momentarily over many hundred ns to a few μs without splitting or destruction of the component occurring, such components would enable a reduced driving outlay in this way. In this case, the dynamically occurring overvoltage spikes are reliably kept below the maximum permissible static voltage of the component since, when the clamping voltage is overshot, the cause of the overvoltage is removed by a lengthened or increased current flow.
Therefore, significantly more freedoms for configuration, in particular with regard to a converter design, would be available to the user (minimized driving outlay and/or minimized outlay with regard to the leakage inductances). What is more, this would be very advantageous for the component manufacturer since the chips could be designed to switch even faster and thus with fewer losses without increasing the risk of excessively high overvoltages during turn-off under extreme conditions (overcurrent or short-circuit case).
In order to achieve this, the design of an IGBT chip should also be devised in a particular manner. Thus, inter alia, the rear side emitter efficiency and the field stop layer are to be dimensioned in a suitable way (see, for example, M. Otsuki et al. “1200 V FS-IGBT module with enhanced clamping capability”; Proceedings of the 2004 ISPSD, pages 339-342, 2004). For these and other reasons, there is a need for the present invention.